SHENOUDA and DAVIDSON: A FRAMEWORK FOR DESIGNING MIMO SYSTEMS WITH DECISION FEEDBACK EQUALIZATION 403
P
sx
y
G
s
DFE
B
Quantizer
^
Transmitt er DFE receiver
Fig. 1. MIMO transceiver using Decision Feedback Equalization.
y
Quantizer
Modulo
G
Tomlinson-Harashima Transmitter Receiver
u
i
P
Modulo
vx
-
B
P
v
x
-
B
(a)
(b)
s
THP
^
Fig. 2. (a) MIMO transceiver with Tomlinson-Harashima precoding (b) Equivalent linear transmitter model for Tomlinson-Harashima precoding system
M
M−1
E{|s
k
|
2
} for all k except the first one [9]. For moderate
to large values of M this power increase can be neglected and
the approximation E{vv
H
} = I is often used; e.g., [5], [10]. If
we assume negligible precoding loss, the average transmitted
power constraint can be written as E
v
{x
H
x} = tr(P
H
P) ≤
P
total
.
The vector of received signals in a TH precoded system can
be written as
y = HPC
−1
u + n, (6)
where n is the vector of additive noise which is assumed to
have zero-mean and a covariance matrix E{nn
H
} = R
n
.At
the receiver, the feedforward processing matrix G is used to
obtain an estimate
ˆ
u = GHPC
−1
u + Gn of the modified
data symbols u. Following this linear receive processing step,
the modulo operation is used to obtain
ˆ
s
THP
by eliminating
the effect of the periodic extension of the constellation caused
by the integer vector i. In terms of the modified data symbols,
the error signal
e =
ˆ
u − u = GHPv + G
H
n − Cv (7)
can be used to define a Mean Square Error matrix
E =E
v
{ee
H
} = CC
H
− CP
H
H
H
G
H
− GHPC
H
+ GHPP
H
H
H
G
H
+ GR
n
G
H
. (8)
Assuming negligible precoding loss and that the vector i is
eliminated by the receiver modulo operation (which occurs
with high probability, even at reasonably low SNRs), the error
signal in (7) is equivalent to
ˆ
s
THP
−s. Hence, the mean square
error matrix, E, of the estimate
ˆ
s
THP
of the TH precoding
model is the same as that of the estimate
ˆ
s
DFE
of the DFE
model under the assumption of correct previous decisions in
the DFE.
C. General Model
From (4) and (8), we observe that the MSE matrix of both
systems can be rewritten as:
E = CC
H
− CP
H
H
H
G
H
− GHPC
H
+ GR
y
G
H
, (9)
where R
y
= HPP
H
H
H
+ R
n
. It can also be observed that
linear transceivers are a special subclass of both system models
with the feedback matrix B = 0 (or, equivalently, C = I); see
Figs 1 and 2. Our objective is to jointly design the matrices
G, C and P according to criteria that are functions of E,
subject to a constraint on the average transmitted power.
III. O
PTIMAL FEEDFORWARD AND FEEDBACK MATRICES
We will consider the joint design of the transceiver matrices
G, C and P so as to optimize system design criteria that
are expressed as (increasing) functions of the (logarithm of
the) MSE of each individual data stream, E
ii
, subject to
the transmitted power constraint tr(P
H
P) ≤ P
total
. We will
adopt a three-step design approach. First, an expression for
the optimal feedforward matrix G will be found as a function
of C and P. Second, using the expression for the optimal G,
an expression for the optimal C will be found as a function
of P. Finally, using the obtained expressions for the optimal
G and C, we will design the optimal precoder P.